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PDF AD604 Data sheet ( Hoja de datos )
| Número de pieza | AD604 | |
| Descripción | Dual/ Ultralow Noise Variable Gain Amplifier | |
| Fabricantes | Analog Devices | |
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FEATURES
Ultralow Input Noise at Maximum Gain:
0.80 nV/√Hz, 3.0 pA/√Hz
Two Independent Linear-in-dB Channels
Absolute Gain Range per Channel Programmable:
0 dB to +48 dB (Preamp Gain = +14 dB), through
+6 dB to +54 dB (Preamp Gain = +20 dB)
؎1.0 dB Gain Accuracy
Bandwidth: 40 MHz (–3 dB)
300 k⍀ Input Resistance
Variable Gain Scaling: 20 dB/V through 40 dB/V
Stable Gain with Temperature and Supply Variations
Single-Ended Unipolar Gain Control
Power Shutdown at Lower End of Gain Control
Can Drive A/D Converters Directly
APPLICATIONS
Ultrasound and Sonar Time-Gain Control
High Performance AGC Systems
Signal Measurement
Dual, Ultralow Noise
Variable Gain Amplifier
AD604
FUNCTIONAL BLOCK DIAGRAM
PAO
–DSX
+DSX
VGN
DIFFERENTIAL
ATTENUATOR
R-1.5R
PAI LADDER NETWORK
0 TO –48.4dB
PROGRAMMABLE
ULTRALOW NOISE
PREAMPLIFIER
G = 14–20dB
PRECISION PASSIVE
INPUT ATTENUATOR
GAIN CONTROL
AND SCALING
VREF
AFA
FIXED GAIN
AMPLIFIER
+34.4dB
OUT
VOCM
PRODUCT DESCRIPTION
The AD604 is an ultralow noise, very accurate, dual channel,
linear-in-dB variable gain amplifier (VGA) optimized for time-
based variable gain control in ultrasound applications; however
it will support any application requiring low noise, wide bandwidth
variable gain control. Each channel of the AD604 provides a
300 kΩ input resistance and unipolar gain control for ease of
use. User determined gain ranges, gain scaling (dB/V) and dc
level shifting of output further optimize application performance.
Each channel of the AD604 utilizes a high performance pre-
amplifier that provides an input referred noise voltage of
0.8 nV/√Hz. The very accurate linear-in-dB response of the
AD604 is achieved with the differential input exponential amplifier
(DSX-AMP) architecture. Each of the DSX-AMPs comprise a
variable attenuator of 0 dB to 48.36 dB followed by a high speed
fixed gain amplifier. The attenuator is based on a seven stage
R-1.5R ladder network. The attenuation between tap points
is 6.908 dB and 48.36 dB for the ladder network.
Each independent channel of the AD604 provides a gain range
of 48 dB which can be optimized for the application by program-
ming the preamplifier with a single external resistor in the
preamp feedback path. The linear-in-dB gain response of the
AD604 can be described by the equation: G (dB) = (Gain
Scaling (dB/V ) × VGN (V )) + (Preamp Gain (dB) – 19 dB).
Preamplifier gains between 5 and 10 (+14 dB and +20 dB)
provide overall gain ranges per channel of 0 dB through +48 dB
and +6 dB through +54 dB. The two channels of the AD604
can be cascaded to provide greater levels of gain range by bypass-
ing the 2nd channel’s preamplifier. However, in multiple channel
systems, cascading the AD604 with other devices in the AD60x
VGA family, which do not include a preamplifier may provide
a more efficient solution. The AD604 provides access to the
output of the preamplifier allowing for external filtering be-
tween the preamplifier and the differential attenuator stage.
The gain control interface provides an input resistance of
approximately 2 MΩ and scale factors from 20 dB/V to
30 dB/V for a VREF input voltage of 2.5 V to 1.67 V respect-
ively. Note that scale factors up to 40 dB/V are achievable
with reduced accuracy for scales above 30 dB/V. The gain scales
linear-in-dB with control voltages of 0.4 V to 2.4 V with the
20 dB/V scale. Below and above this gain control range, the gain
begins to deviate from the ideal linear-in-dB control law. The
gain control region below 0.1 V is not used for gain control. In
fact when the gain control voltage is <50 mV the amplifier
channel is powered down to 1.9 mA.
The AD604 is available in a 24-pin plastic SSOP, SOIC and DIP,
and is guaranteed for operation over the –40°C to +85°C
temperature range.
REV. 0
Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.
One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700 World Wide Web Site: http://www.analog.com
Fax: 617/326-8703
© Analog Devices, Inc., 1996
1 page  
Typical Performance Characteristics (per Channel)–AD604
(Unless otherwise noted G (preamp) = +14 dB, VREF = 2.5 V (20 dB/V Scaling), f = 1 MHz, RL = 500 ⍀, CL = 5 pF, TA = +25؇C, VSS = ؎5 V)
50
40
3 CURVES
30 –40°C,
+25°C,
+85°C
20
10
0
–10
0.1 0.5 0.9 1.3 1.7 2.1 2.5 2.9
VGN – Volts
60
G (PREAMP) = +14dB
50 (0dB – +48dB)
40
G (PREAMP) = +20dB
30 (+6dB – +54dB)
20
10
0 DSX ONLY
(–14dB – +34dB)
–10
–20
0.1 0.5 0.9 1.3 1.7 2.1 2.5 2.9
VGN – Volts
50
40
30dB/V
VREF = 1.67V
30
ACTUAL
ACTUAL
20
20dB/V
VREF = 2.50V
10
0
–10
0.1 0.5 0.9 1.3 1.7 2.1 2.5 2.9
VGN – Volts
Figure 1. Gain vs. VGN
Figure 2. Gain vs. VGN for Different Figure 3. Gain vs. VGN for Different
Preamp Gains
Gain Scalings
40
37.5
THEORETICAL
35
32.5
ACTUAL
30
27.5
25
22.5
20
1.25 1.5 1.75
2
VREF – Volts
2.25
2.5
Figure 4. Gain Scaling vs. VREF
2.0
1.5
1.0
0.5 –40°C +25°C
0
–0.5 +85°C
–1.0
–1.5
–2.0
0.2 0.7 1.2 1.7 2.2 2.7
VGN – Volts
Figure 5. Gain Error vs. VGN at
Different Temperatures
2.0
1.5
1.0
0.5 FREQ = 1MHz
0
–0.5
–1.0
–1.5
FREQ = 10MHz
FREQ = 5MHz
–2.0
0.2 0.7 1.2 1.7 2.2 2.7
VGN – Volts
Figure 6. Gain Error vs. VGN at
Different Frequencies
2.0
1.5
1.0
0.5
0
–0.5
–1.0
–1.5
–2.0
0.2
20dB/V
VREF = 2.50V
30dB/V
VREF = 1.67V
0.7 1.2 1.7 2.2 2.7
VGN – Volts
Figure 7. Gain Error vs. VGN for
Different Gain Scalings
25 25
N = 50
N = 50
VGN1 = 1.0V
VGN1 = 2.50V
20
VGN2 = 1.0V
20
VGN2 = 2.50V
∆G(dB) =
∆G(dB) =
G(CH1) – G(CH2)
G(CH1) – G(CH2)
15 15
10 10
55
0
–1.0 –0.8 –0.6 –0.4 –0.2 0.1 0.3 0.5 0.7 0.9
DELTA GAIN – dB
0
–1.0 –0.8 –0.6 –0.4 –0.2 0.1 0.3 0.5 0.7 0.9
DELTA GAIN – dB
Figure 8. Gain Match; VGN1 = VGN2 = Figure 9. Gain Match: VGN1 = VGN2 =
1.0 V
2.50 V
REV. 0
–5–
5 Page 
AD604
1 –DSX1
VGN1 24
2 +DSX1
VREF 23
3 PAO1
OUT1 22
4 FBK1
GND1 21
5 PAI1
VPOS 20
6 COM1
VNEG 19
AD604
7 COM2
VNEG 18
8 PAI2
VPOS 17
9 FBK2
GND2 16
10 PAO2
OUT2 15
11 +DSX2
VOCM 14
12 –DSX2
VGN2 13
Figure 38. Shutdown of Preamplifiers Only
Differential Ladder (Attenuator)
The attenuator before the fixed gain amplifier of the DSX is
realized by a differential seven-stage R-1.5R resistive ladder net-
work with an untrimmed input resistance of 175 Ω single-ended
or 350 Ω differentially. The signal applied at the input of the
ladder network (Figure 39) is attenuated by 6.908 dB per tap;
thus, the attenuation at the first tap is 0 dB, at the second,
13.816 dB, and so on, all the way to the last tap where the
attenuation is 48.356 dB. A unique circuit technique is used to
interpolate continuously between the tap points, thereby provid-
ing continuous attenuation from 0 to –48.36 dB. You can think
of the ladder network together with the interpolation mechanism
as a voltage-controlled potentiometer.
Since the DSX is a single-supply circuit, some means of biasing
its inputs must be provided. Node MID together with the
VOCM buffer performs this function. Without internal biasing,
the user would have had to dc bias the inputs externally. If not
done carefully, the biasing network can introduce additional
noise and offsets. By providing internal biasing, the user is
relieved of this task and only needs to ac couple the signal into
the DSX. It should be made clear again that the input to the
DSX is still fully differential if driven differentially, i.e., pins
+DSX and –DSX see the same signal but with opposite polarity
(see Differential Input VGA Application). What changes is the
load as seen by the driver; it is 175 Ω when each input is driven
single ended, but 350 Ω when driven differentially. This can be
easily explained when thinking of the ladder network as just two
175 Ω resistors connected back-to-back with the middle node,
MID, being biased by the VOCM buffer. A differential signal
applied between nodes +DSX and –DSX will result in zero cur-
rent into node MID, but a single-ended signal applied to either
input +DSX or –DSX while the other input is ac grounded, will
cause the current delivered by the source to flow into the
VOCM buffer via node MID.
The ladder resistor value of 175 Ω was chosen to provide the
optimum balance between the load driving capability of the
preamplifier and the noise contribution of the resistors. One fea-
ture of the X-AMP architecture is that the output referred noise
is constant versus gain over most of the gain range. This can be
easily explained by looking at Figure 39 and observing that the
tap resistance is equal for all taps after only a few taps away
from the inputs. The resistance seen looking into each tap is
54.4 Ω which makes 0.95 nV/√Hz of Johnson noise spectral
density. Since there are two attenuators, the overall noise con-
tribution of the ladder network is √2 times 0.95 nV/√Hz or
1.34 nV/√Hz, a large fraction of the total DSX noise. The rest
of the DSX circuit components contribute another 1.20 nV/√Hz
which together with the attenuator produces 1.8 nV/√Hz of
total DSX input referred noise.
AC Coupling
As already mentioned, the DSX portion of the AD604 is a
single-supply circuit and therefore its inputs need to be ac
coupled to accommodate ground-based signals. External
capacitors C1 and C2 in Figure 35 level shift the ground refer-
enced preamplifier output from ground to the dc value estab-
lished by VOCM (nominal 2.5 V). C1 and C2, together with
the 175 Ω looking into each of DSX inputs (+DSX and –DSX),
will act as high pass filters with corner frequencies depending on
the values chosen for C1 and C2. For example, if C1 and C2
are 0.1 µF, then together with the 175 Ω input resistance seen
into each side of the differential ladder of the DSX, a –3 dB high
pass corner at 9.1 kHz is formed.
If the AD604 output needs to be ground referenced, then an-
other ac coupling capacitor will be required for level shifting.
This capacitor will also eliminate any dc offsets contributed by
the DSX. With a nominal load of 500 Ω and a 0.1 µF coupling
capacitor, this adds a high pass filter with –3 dB corner fre-
quency at about 3.2 kHz.
The choice for all three of these coupling capacitors depends on
the application. They should allow the signals of interest to pass
unattenuated, while at the same time they can be used to limit
the low frequency noise in the system.
+DSX
R –6.908dB R –13.82dB R –20.72dB R –27.63dB R –34.54dB R –41.45dB R –48.36dB
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
1.5R
MID
1.5R
R
–DSX
NOTE: R = 96Ω
1.5R = 144Ω
R
1.5R
R
1.5R
R
1.5R
R
1.5R
R
1.5R
R
1.5R
175Ω
175Ω
Figure 39. R–1.5R Dual Ladder Network.
REV. 0
–11–
11 Page | ||
| Páginas | Total 20 Páginas | |
| PDF Descargar | [ Datasheet AD604.PDF ] | |
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